有人试过用开关电源给胆机供电吗

denggboo

The inductance of L2 is chosen so that current continues to flow for most of the "off" period at full load. You can see this effect in the SPICE simulation (Fig.3). As previously described, the use of an LC filter ensures that the output voltage depends on the duty cycle, as required for PWM control.

Diodes D3 & D4 have to withstand a reverse voltage of about 900V during the transistor "on" period, as well as some voltage spikes passed from the primary to the secondary by inter-winding capacitance. Note that these spikes are generated during the "off" period by primary leakage inductance - they do not transform to the secondary inductively. Hence, the BYV26G fast avalanche diode with a peak reverse voltage of 1400V was chosen for the job. These are available locally from RS Components (Cat. 216-9397).

Diodes D1 & D2 provide a low impedance return path for inductor (L2) current during the switch-off period. They also combine in the D2-C2-C1-D3 and D4-C2-C1-D1 circuits to clamp the secondary voltage to 盫OUT.

One of the advantages of this clamping method is that it passes much of the energy stored in the core of T1 to the load. This energy would otherwise recirculate through the primary side protection diodes (D8 & D9), as well as dissipate in a more aggressive clamp or snubber network with higher losses.

At power up, the clamp forms a short circuit across the secondary until C1 & C2 are charged, so 100 resistors have been inserted to limit the maximum current. The clamp is important in reducing the inductive kick of the primary winding (as opposed to the primary leakage inductance whose kick can not be avoided). The effect of the secondary clamping can be seen as a plateau during the "off" period in the SPICE simulation and the measured primary voltages of Fig.3 and Fig.4(a), respectively. This waveform resembles a square wave at any duty cycle.

An important parameter in the design of the power sections of the circuit is the choice of the secondary voltage to output voltage differential. This is needed to provide headroom to compensate for a drop in the secondary voltage with increased power output, the difference being made up by the by duty cycle variation controlled by the TL494.

Secondary voltage drop has several sources: ohmic losses in inductor coils, non-linearity of the cores, 100Hz ripple due to discharge of mains storage capacitors C6 & C7 and voltage drop across C8 which is charged and discharged every switching cycle. The latter two effects contribute to a primary voltage ripple of 22V and 9V peak-to-peak respectively at full output power, which manifests as a 63V ripple across the secondary. Choosing a 100V differential allows output voltages of up 800V to be delivered by this supply at full power and regulation.

denggboo

The output voltage is smoothed by a capacitive divider consisting of two 10μF capacitors (C1 & C2) rated at 450V. Alternatively, the 400V version has a higher output current and so 47μF capacitors rated at 350V (or higher) should be used.

At this rating, they are each available in a small package which is easily accommodated in the space provided on the PC board for the original 5V supply components. Their capacitance contributes only about 700mV of 60kHz ripple (0.1%) at full load.

Two 180kΩ resistors across the output set a minimum load current, ensuring that the PWM controller switches Q1 & Q2 on for at least a small portion of each period. Resistor chain R3-R6 divides down the high-voltage output, developing a lower voltage feedback signal that is applied to the non-inverting error amplifier input of the TL494.

Output voltage regulation is achieved by varying the duty cycle so that the voltage applied to the TL494’s non-inverting amplifier input (pin 1) equals the voltage on the inverting input (pin 2). In this case, 2.5V is applied to pin 2 via resistive divider R7 & R8. Hence, if R6=4.7kΩ, then R3 + R4 + R5 should equal 1311kΩ for 700V output, or 747kΩ for 400V.

The filament supply is provided by a simple modified version of the original 12V secondary. Unfortunately, we can’t use the TL494 to regulate the 12V supply because the original circuit used a coupled inductor shared between the secondaries for this purpose. Our two secondaries now have a high voltage between them. Hence, an LM350T adjustable 3A regulator is used to derive the 12.7V supply. It also powers the 12V cooling fan and must be fitted with a heatsink.

This secondary also supplies power to the TL494 via D7 & C5, as in the original circuit. If a 24V filament supply is required, the more common 7824 1A regulator can be used, as less current is required. The cooling fan can be run from 24V using a 47Ω 5W series dropping resistor.

When the HV supply is only lightly loaded, the duty cycle is so small that the filament supply is not able to deliver its rated current. This can occur at power-on because no plate current flows when the filaments are cold. However, without HV current the filaments can not warm up. To avoid this stalemate, an auxiliary voltage control circuit consisting of resistors R13 & R14 and diode D12 is employed.

During normal operation, D12 is reverse-biased and the voltage at pin 1 of the TL494 is derived from the HV supply alone. However, when the filament voltage drops, the cathode of D12 becomes less positive until, at about 1.9V, the diode conducts and prevents the filament voltage dropping any further. With the resistor values shown, this threshold is set at about 10.5V.


This photo and the photo to the right show how to wind and insulate one layer of the HV secondary. The layer must start and finish on the secondary face of the former, adjacent to the PC board pins. Starting on the pin end of the former, close-wind one complete layer (no overlaps). After the layer is complete, apply about 1 and 1/4 turns of high-voltage insulation tape. Position the start of the tape approximately as shown.
Return the free end of the wire back to the start side, and then bind over it with the end of the insulation tape. The aim is to insulate the return wire from the layer beneath and the one to follow. With the return wire sealed between the two layers of tape, continue winding the next layer, or terminate at the pins if it's the final layer.
A view of the completed trans-former. Note how the centre-tapped connection to the final (12V) winding exits through a small hole in the tape, rather than being terminated at the pins.

denggboo

Circuit protection

Care is required to ensure that the deadtime control circuit connected to pin 4 of the TL494 operates correctly in the modified circuit. The function of the deadtime control is to provide over-voltage and over-current protection if the transformer core saturates.

Primary side current is sensed using T3, a small current transformer. Its primary winding is connected in series with the primary of the main switching transformer. T3 employs a very large transformation ratio (n of about 180), combined with a relatively small resistance across its secondary winding.

This resistance swamps the effects of primary inductance, such that the voltage drop across the transformer is due only to the resistance. The secondary voltage is then proportional to the primary current at about 2V per amp with n = 180 and R9 = 350Ω.

The resultant signal is rectified by D10, smoothed by C10 and applied to potential divider R10 & R11. When the voltage at the midpoint of this divider exceeds about 0.6V, Q3 conducts and a positive voltage is applied to pin 4 of the TL494 through diode D11. A voltage of 0V on this pin sets a minimum deadtime of 4% while at 3.3V, Q1 and Q2 do not turn on at all.

The values shown for R10 & R11 set a threshold current of about 2.8A but you could vary this by altering these resistors. Transistor Q7 and LED1 were added to the circuit to indicate activation of the over-current protection.

Transistor Q4 and the voltage divider connected to its base provides protection against output voltage imbalances by injecting current into the base of Q3 under fault conditions. The voltage divider in the original circuit was designed to produce about 0V at the base of Q4 under normal conditions. However, since the modified supply no longer generates the negative voltages of the original circuit but still has the +12V circuit, this would upset the current balance in the resistor network. Catastrophic failures aside, output voltage regulation prevents over-voltage anyway, so the easiest solution is to disable this part of the circuit by removing the associated components (shown shaded in green on the circuit diagram).

Your circuit may use a different configuration to the one shown here. For example, the LM339 comparator is frequently used for over-voltage detection. If the voltage on pin 4 of the TL494 exceeds about 0.3V under no load, simply disconnect any resistor running from the 12V supply to the control circuitry connected to this pin.

Note that some power supplies do not use discrete components in the protection circuitry at all. Unfortunately, this article can not hope to cover all possible variations. If you do not feel confident in modifying the existing circuitry, then it is recommended that you construct the circuit shown in Fig.2 and use it to replace the protection circuits connected to pin 4.


A view of the reassembled PC board showing the newly rewound transformer (T1), HV filter inductor (L2) and HV capacitors (C1 & C2). L2 can be secured to the board using non-acidic silicone sealant.Selecting component values

Of major importance in this design is the correct selection of filter inductor L2. If the inductance of L2 is too small, the circuit reduces to a standard capacitive voltage doubler configuration and the dependence of output voltage on duty cycle is lost. Alternatively, if it is too large, the voltage developed across it each half-cycle is insufficient to raise the current required by the load.

In practice, the optimum value is about 450μH for a 700V output (150μH for 400V).

Another challenge is choosing appropriate values for the primary and secondary damper networks – R2 & C4 and R1 & C3. The former is needed to damp the leakage inductance component of the primary, which exists in all coils due to a small amount of primary flux that’s not coupled to the secondary. The energy stored in this flux during the "on" period (1/2LI2) generates a current which charges the transistor output capacitance and transformer stray capacitance (C0) when the transistors turn off. In the absence of resistive losses, this energy is fully transferred into capacitive energy (1/2CV2).

If the transformer is rewound as described in the construction section, it will have a leakage inductance of about 10μH. As this is less than 1% of the 3.5mH total primary inductance, it is quite acceptable. C0 is about 270pF, while at full 150W load, the primary current can reach 1.7A.

Plugging in the values gives a voltage spike of about 350V. This adds to the voltage drop across the primary inductance during the "on" period and can destroy the output transistors if the ringing is not damped (protective diodes D8 & D9 offer limited protection due to their finite resistance and turn-on time).

The damper network has the side effect of dissipating energy not only when the transistors switch off but also when they turn on. Making the damper capacitor too large leads to the energy dissipation at turn-on far outstripping the parasitic energy.

The parasitic energy is just the energy stored in the leakage inductance and equates to a power of 0.9W at full load. We’ve selected a 2W resistor for R2, which leaves 1.1W to be dissipated during switch-on. A capacitor of 1nF will dissipate about 1W in R2 during the "on" period.

R2 should be 50Ω for critical damping. Making R2 smaller does not increase damping; rather, the damper capacitor effectively acts in parallel with the primary winding to change its ringing frequency.

A second source of ringing occurs when current through L2 drops to zero during the "off" period. Depending
on the polarity of the half-period, either diode D1 or D2 stop conducting. However, the voltage across L2 can not change instantaneously, due to the energy stored in the diode and switching transistor output capacitance. The resultant ringing is dissipated by the damper network across the secondary and by hysteresis losses in L2.

Construction

You should read the earlier SILICON CHIP article (October 2003) on modifying a PC power supply prior to commencing construction. Note that quite a few more components need to be removed here, since most of the secondary side is unsuitable for HV operation.

Begin by removing the large low-voltage secondary capacitors and inductors. The 5V rectifier and associated heatsink also need to be removed, as well as the secondary RC damping network. The multiple power supply leads for the various output voltages are best unsoldered and removed using a large (60W) soldering iron.

Care must be taken when desoldering the ferrite transformer (T1) to avoid melting the former plastic and loosening the pins. Remove all resistors and links leading from the 5V supply to the control circuitry of the TL494.


Fig.3: the output from a SPICE simulation of transformer primary voltage and toroid current waveforms. The simulation results closely follow the actual waveforms measured in the working prototype.Transformer preparation

denggboo

The next job is to remove the existing windings from the ferrite transformer in preparation for the rewind. Begin by carefully removing the tape binding the core sections together, as it can be reused later. Soak the transformer in methylene chloride paint stripper overnight to remove most of sealing varnish. Note that gloves and protective eyeglasses should be worn when working with paint stripper.

If you don’t wish to wait overnight, then the transformer can be warmed prior to dipping for a few minutes with a hair drier held at close range. After about 15 minutes, the transformer can be gently removed and light pressure applied by means of a screwdriver between the slab section of the core and the former, allowing the latter to be released.

It is advisable to remove the E-section out of the former immediately by pressing gently on the centre prong of the "E" (the outer prongs are fragile and easily broken). Care needs to be taken here since this is the only component that is not easily replaced.

If the E-section won’t separate with light pressure, then wash the transformer thoroughly and use a razor blade and sidecutters to slice and remove sections of insulation and copper wire to free it up. Complete the transformer disassembly by washing all components and removing all the wire and insulation from the former.

Transformer rewind

Great care must be taken with the transformer rewind to ensure primary to secondary isolation. In particular, make sure that each layer is completely covered with the tape, right up to the shoulders of the former, so that turns from different layers can not touch.

Except where noted, there should be no gaps between the start and finish of a layer and the shoulders of the former. This ensures that wire from the next layer can not creep into the gap and potentially make contact with the preceding layer.

The HV secondary winding goes on first, using 28 SWG (0.4mm) enamelled copper wire. Three layers are required, producing 117 turns in total. For the 400V version, use three layers of 24 SWG wire instead, producing 75 turns in total. It does not matter if your winding is a few turns short. The inner layer is the "hotter" end of the winding. It must be connected to the third pin from the edge of the PC board on the secondary side of the former.

A layer of polyester high-voltage tape is used to insulate each layer. Suitable "3M" brand high-voltage polyester tape is available from Farnell (cat. no. 753-002). Note that this tape is 19mm wide, whereas the standard former requires 17mm tape. To obtain the correct width, stick strips about 10cm long onto a clean plastic surface (such as transparency film) and trim off 2mm using a razor blade and straight edge.

One end of the tape is placed over the top of a completed layer and the free end of the wire is returned over the top and sealed by one turn of the tape (see photos). The wire must be returned on the pin face rather than the sides of the former otherwise there will not be sufficient room for the core.

The copper strip used in the original transformer to reduce inter-winding capacitance is not needed here because the windings are not interleaved. With all three layers in place, insulate the HV secondary with three layers of polyester high-voltage tape.

Using the same technique, the primary is now wound in two layers with 24 SWG wire, for a total of 40 turns. Note that the first layer will be 25 turns, whereas the second layer will be only 15 turns. This leaves a gap between the finish of the winding and the shoulder of the former. Before applying the inter-winding insulation over the second layer, this gap must be filled in with tape.

To achieve this, cut strips of high-voltage tape of the appropriate width and build up the gap to the same height as the windings. The idea here is to achieve a smooth, level surface for the final winding. That done, insulate the primary with two layers of high-voltage tape.

Finally, the 12V secondary is wound with 12 centre-tapped turns in a single layer using 24 SWG wire and insulated with a single layer of high-voltage tape. It’s easier if the centre-tap connection is not terminated at the pins (see photo).

The transformer core can now be fitted, making sure that the abutting faces are perfectly clean. This is necessary because the ferrite core is of very high permeability material (ie, μe about 2000). An air gap of only one two thousandths of the core length (about 25 microns) will be sufficient to halve the coil’s inductance. The core sections are pressed together tightly, bound with the original tape, and the whole assembly sprayed with lacquer and left a few hours to dry.

It is best if the former pins are masked with tape prior to spraying to make subsequent soldering easier.


Fig.4(a): this scope waveform was measured across the primary of transformer T1 and shows the alternate switching of transistors Q1 and Q2. Notice how the secondary voltage clamp has flattened the peaks of the waveform to produce a square-wave voltage that's independent of the duty cycle. Note also that the waveform peaks are slanted slightly due to the discharging of C8.
Fig.4(b): the voltage across toroid L2 over several cycles. The peaks of about 370V occur during the "off" period when L2 discharges into the smoothing capacitors (C1 & C2). Some ringing occurs when the current drops to zero, as described in the text. During the "on" period, the voltage across L2 equals the difference between the secondary and output voltages, decreasing steadily as C8 is charged.
Fig.4(c): 60kHz output ripple at full load is about 2V p-p at 700V DC.
Fig.4(d): 100Hz hum can be seen on top of the 60kHz ripple and amounts to only about 0.6Vp-p.Note: for safety reasons, these waveforms were all taken with the SMPS connected to the mains via an isolation transformer. Don't attempt this unless you know exactly what you are doing.

Toroid rewind

Toroid L2 is wound next. A key requirements for L2 are that its insulation should withstand about 500V and it must be able to dissipate the heat generated by hysteresis in its core. The latter is not to be confused with ohmic losses in the windings (which are small here) and arises because the core does not demagnetise at zero current. This remnant magnetism is removed by reverse current every cycle and manifests itself as heat. In practice, hysteresis losses can be reduced by using a larger core size for a given value of inductance.

With this in mind, L2 consists of two standard 25 x 10mm toroids glued at the faces to form a single core. This gives the required inductance in a single layer and reduces hysteresis heating. Suitable core material is the standard yellow/white or green/yellow ferrite typically used in PC power supplies.

The original low-voltage windings are discarded and the faces of the toroids thoroughly cleaned before glu-
ing. L2 is wound in a single layer with 56 turns of 24 SWG wire. For 400V versions, use 33 turns of 24 SWG wire.

In operation, the core should only get warm to the touch at full power (make sure you turn off the converter before checking this!).

PC board rebuild

denggboo

All of the necessary components can now be installed on the PC board.

The HV section occupies the area previously taken up by the 5V supply. Although the exact PC board layout varies between manufacturers, a typical design allows easy accommodation of all the components shown in Fig.1. However, you might need to break a few copper tracks with a sharp knife or engraver tool and add a few links with insulated hook-up wire.

Important: be sure to leave at least 1mm clearance between all high-voltage tracks in this part of the circuit.

Remember to install the resistive divider from the HV supply to pin 1 of the TL494 in place of the original divider running from the 5V supply. Inductor L4 and capacitor C9 are mounted directly across the rear of the output terminals (see photos). The inductor consists of 5-6 turns of hookup wire wound around a small toroid.

The 12V circuit occupies its previous location but make sure that all components not shown in Fig.1 are removed. The 4μH inductor (L3) can be salvaged from the original 5V supply and consists of 7-8 turns of copper wire around a ferrite rod.

Once you are certain that no HV is fed anywhere except as shown in Fig.1, you are ready to apply power to the circuit. It would be useful to have a load available to check operation at reasonable power levels. I used several strips of five 4.7kΩ 5W resistors connected in series to provide a 25W per strip load at 700V.

Warning: switchmode power supplies have been known to explode on failure, expelling particles of component material such as metal, epoxy and glass at high speed. Close the case or wear protective eyeglasses before applying power!

If the circuit fails to deliver substantial power, the problem might be due to the current protection circuit. Check that the voltage on pin 4 of the TL494 does not exceed about 0.3V under normal load. If it does, this part of the circuit is malfunctioning. Follow the techniques described in the circuit protection section above to track down the problem.

Finally, because the modified converter is less efficient than the original, it requires better cooling when operating at full power. This can be achieved by switching the cooling fan around so that it forces the air into the case.

WARNING!
This is NOT a project for the inexperienced. Do not even think of opening the case of a PC switchmode power supply (SMPS) unless you have experience with the design or servicing of such devices or related high-voltage equipment.

Some of the SMPS circuitry is at full mains potential. In addition, the high-voltage DC output from this supply could easily kill you. Beware of any residual charge on the mains and output capacitors, even if turned off for some time.

The metal case and ground (0V) outputs of all PC power supplies are connected to mains earth. You should verify that these connections are in place after completing any modifications; under no circumstances should the output be floated!

DO NOT attempt to modify a SMPS unless you are fully competent and confident to do so.

denggboo

图片1[localimg=680,426]1[/localimg]

denggboo

继续

徐州老史

原帖由 denggboo 于 2008-2-18 14:20 发表
All of the necessary components can now be installed on the PC board.

The HV section occupies the area previously taken up by the 5V supply. Although the exact PC board layout varies between m ...

denggboo

也不董啊,期待高人翻译

大.非

谁能把字母变成汉字？？？？？？？？？

jjxa